Subranging analog to digital converter with multi-phase clock timing

ABSTRACT

An N-bit analog to digital converter includes a reference ladder, a track-and-hold amplifier connected to an input voltage, a coarse ADC amplifier connected to a coarse capacitor at its input and having a coarse ADC reset switch controlled by a first clock phase of a two-phase clock, a fine ADC amplifier connected to a fine capacitor at its input and having a fine ADC reset switch controlled by a second clock phase of the two-phase clock, a switch matrix that selects a voltage subrange from the reference ladder for use by the fine ADC amplifier based on an output of the coarse ADC amplifier, and wherein the coarse capacitor is charged to a coarse reference ladder voltage during the first clock phase and connected to the T/H output during the second clock phase, wherein the fine capacitor is connected to a voltage subrange during the first clock phase and to the T/H output during the second clock phase, and an encoder that converts outputs of the coarse and fine ADC amplifiers to an N-bit output.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a Continuation-in-Part of application Ser. No. 10/153,709, Filed: May 24, 2002, Titled: DISTRIBUTED AVERAGING ANALOG TO DIGITAL CONVERTER TOPOLOGY, Inventors: MULDER et al.; is a continuation of application Ser. No. 10/158,773, filed on May 31, 2002, Titled: SUBRANGING ANALOG TO DIGITAL CONVERTER WITH MULTI-PHASE CLOCK TIMING; and is related to application Ser. No. 10/158,774, Filed: May 31, 2002; Titled: ANALOG TO DIGITAL CONVERTER WITH INTERPOLATION OF REFERENCE LADDER, Inventors: MULDER et al.; application Ser. No. 10/158,595, Filed: May 31, 2002, Titled: HIGH SPEED ANALOG TO DIGITAL CONVERTER, Inventor: Jan MULDER; and application Ser. No. 10/158,193, Filed: May 31, 2002, Inventor: Jan MULDER; Titled: CLASS AB DIGITAL TO ANALOG CONVERTER/LINE DRIVER, Inventors: Jan MULDER et al., all of which are incorporated by reference herein.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to analog to digital converters ADC, and more particularly, to analog to digital converters utilizing track-and-hold amplifiers for high speed operation.

2. Related Art

A subranging analog to digital converter (ADC) architecture is suitable for implementing high-performance ADC's (i.e. high speed, low power, low area, high resolution). FIG. 1 shows the generic two-step subranging architecture, comprising a reference ladder 104, a coarse ADC 102, a switching matrix 103, a fine ADC 105, coarse comparators (latches) 107, fine comparators (latches) 108 and an encoder 106. In most cases, a track-and-hold 101 is used in front of the ADC. In this architecture, an input voltage is first quantized by the coarse ADC 102. The coarse ADC 102 and the coarse comparators 107 compare the input voltage against all the reference voltages, or against a subset of the reference voltages that is uniformly distributed across the whole range of reference voltages. Based on a coarse quantization, the switching matrix 103 connects the fine ADC 105 and the fine comparators 108 to a subset of the reference voltages (called a “subrange”) that is centered around the input signal voltage. The coarse and fine comparators 107, 108 latch the outputs of the coarse and fine ADC's 102, 105 prior to inputting them to the encoder 106.

High-speed high-resolution ADC's usually use a track-and-hold (T/H) or a sample-and-hold (S/H) preceding the ADC. The main distinction between a S/H and a T/H is that a S/H holds the sampled input signal for (almost) a full clock period, whereas a T/H holds the sampled input signal for (almost) half a clock period.

In general, a S/H requires more area and power than a T/H to obtain the same performance. However, the disadvantage of a T/H is that the sampled input signal is available to the ADC for only half a clock period.

Other subranging ADC's are known that can use a T/H instead of a S/H. However, the timing proposed in conventional art has important disadvantages.

Typically, both the coarse and fine ADC amplifiers reset to the T/H output voltage. This leaves much less time available for the coarse ADC amplifiers to amplify the signals and the coarse comparators to decide on a voltage to latch. This will impact a maximum sampling speed F_(sample) that the ADC can run at.

Some ADC's use a T/H, where the same physical circuits are used for performing both the coarse and the fine quantization. This leaves only ¼ of a clock cycle available for performing the coarse quantization, or two time-interleaved sub-ADC's have to be used. This impacts either maximum possible operating speed, or doubles required area and power.

Thus, one of the bottlenecks in subranging ADC's is the limited amount of time available for performing the coarse quantization. Several different timing methods for subranging ADC's are known for optimizing this bottleneck. Unfortunately, most of these solutions require the use of a S/H, or use time-interleaved ADC's. This disadvantageously affects the required power and area.

SUMMARY OF THE INVENTION

The present invention is directed to an analog to digital converter that substantially obviates one or more of the problems and disadvantages of the related art.

There is provided an N-bit analog to digital converter including a reference ladder, a track-and-hold amplifier connected to an input voltage, and a coarse ADC amplifier connected to a coarse capacitor at its input and having a coarse ADC reset switch controlled by a first clock phase of a two-phase clock. A fine ADC amplifier connected to a fine capacitor at its input and has a fine ADC reset switch controlled by a second clock phase of the two-phase clock. A switch matrix selects a voltage subrange from the reference ladder for use by the fine ADC amplifier based on an output of the coarse ADC amplifier. The coarse capacitor is charged to a coarse reference ladder voltage during the first clock phase and connected to the T/H output voltage during the second clock phase, wherein the fine capacitor is connected to a voltage subrange during the first clock phase and to the T/H output voltage during the second clock phase. An encoder converts outputs of the coarse and fine ADC amplifiers to an N-bit output.

In another aspect of the present invention there is provided an N-bit analog to digital converter including a reference ladder, a track-and-hold amplifier tracking an input voltage, a two-phase clock having phases φ₁ and φ₂, and a plurality of coarse ADC amplifiers each connected to a corresponding coarse capacitor at its input. The coarse ADC amplifiers are reset on φ₁ and their corresponding coarse capacitors are connected to the T/H output voltage on φ₂. A plurality of fine ADC amplifiers are each connected to a corresponding fine capacitor at their input. The fine ADC amplifiers are reset on φ₂ and their corresponding fine capacitors are charged to the T/H output voltage on φ₂. A switch matrix selects a voltage subrange from the reference ladder based on outputs of the coarse ADC amplifiers for input to the fine ADC amplifiers on φ₁. An encoder converts outputs of the coarse and fine ADC amplifiers to an N-bit output.

In another aspect of the present invention there is provided a N-bit analog to digital converter including a reference ladder, a track-and-hold amplifier tracking an input voltage, a two-phase clock having phases φ₁ and φ₂, a coarse capacitor connected to the track-and-hold amplifier on φ₂ and to the reference ladder on φ₁, a coarse ADC amplifier that resets on φ₁ and amplifies a voltage on the coarse capacitor on φ₂, and a coarse comparator for latching an output of the coarse ADC amplifier on φ_(1+1 cycle). A fine capacitor is connected to the track-and-hold on φ₂ and to a fine voltage tap of the reference ladder on φ₁, the fine voltage tap selected based on the output of the coarse ADC amplifier. A fine ADC amplifier includes a plurality of cascaded amplifier stages. A first cascaded amplifier stage resets on φ₂ and amplifies a voltage on the fine capacitor on φ_(1+1 cycle), a second cascaded amplifier stage resets on φ_(1+1 cycle) and amplifies the voltage on the fine capacitor on φ_(2+1 cycle), a third cascaded amplifier stage resets on φ_(2+1 cycle) and amplifies the voltage on the fine capacitor on φ_(1+2 cycles), and so on. A fine comparator latches an output of a last cascaded amplifier stage on φ_(1+3 cycles), and an encoder converts outputs of the coarse and fine comparators to an N-bit output.

In another aspect of the present invention there is provided an N-bit analog to digital converter including a reference ladder, a track-and-hold amplifier tracking an input voltage, a two-phase clock having alternating phases φ₁ and φ₂, a plurality of coarse capacitors connected to an output of the track-and-hold on φ₂ and to corresponding coarse taps of the reference ladder on φ₁, and a plurality of coarse ADC amplifiers that reset on φ₁ and amplify voltages on the coarse capacitors on φ₂. A plurality of coarse comparators latches outputs of the coarse ADC amplifiers. A plurality of fine capacitors connected to the output of track-and-hold amplifier on φ₂ and connected to fine voltage taps of the reference ladder on φ₁, the fine voltage taps are selected based on the outputs of the coarse ADC amplifiers. A plurality of fine ADC amplifiers, each including a plurality of cascaded amplifier stages. The cascaded amplifier stages reset and amplify on alternating phases φ₁ and φ₂, wherein amplifiers of the first stage are reset on φ₂ and amplify voltages of the fine capacitors on φ₁, a plurality of fine comparators for latching outputs of a last amplifier stage. An encoder converts outputs of the coarse and fine comparators to an N-bit output.

Additional features and advantages of the invention will be set forth in the description which follows, and in part will be apparent from the description, or may be learned by practice of the invention. The advantages of the invention will be realized and attained by the structure particularly pointed out in the written description and claims hereof as well as the appended drawings.

It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are intended to provide further explanation of the invention as claimed.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention. In the drawings:

FIG. 1 illustrates a conventional subranging ADC architecture;

FIG. 2 illustrates connections of one set of amplifiers of the present invention;

FIG. 3 illustrates a timing diagram for operation of the circuit of FIG. 2;

FIG. 4 shows a pipelined timing diagram for the present invention;

FIG. 5 shows the timing diagram of the present invention in additional detail; and

FIGS. 6 and 7 show a flow chart for the timing diagram of FIG. 5; and

FIGS. 8 and 9 show the coarse and fine ADC's as amplifier arrays.

FIG. 10 shows the circuit of FIG. 2 with FET transistors used as switches.

FIG. 11 shows cascaded coarse and fine amplifier stages.

DETAILED DESCRIPTION OF THE INVENTION

Reference will now be made in detail to the preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings.

This disclosure describes a subranging ADC that uses a two-phase clock timing method that permits the use of a T/H instead of a S/H, thus enabling a low-power, low-area implementation on a chip. The timing technique described herein can use a T/H, instead of a S/H, and does not require time-interleaved ADC's in order to realize high-speed operation.

FIG. 2 shows a single coarse amplifier A_(C) and a single fine amplifier A_(F) that illustrate the proposed timing method of the present invention. Preferably, the coarse amplifier A_(C) and the fine amplifier A_(F) are implemented using auto-zero amplifiers. See, e.g., http://www.web-ee.com/primers/files/auto-zero_amps.pdf, for a general discussion of auto-zero amplifiers. (To the extent the overall block diagram of the architecture is the same as that in FIG. 1, the same reference labels will be used herein. It is also understood that the coarse ADC 102 and the fine ADC 105 actually include an array of coarse amplifiers and an array of fine amplifiers. See, e.g., The Circuits and Filters Handbook, Wai-Kai Chen, ed., 1995, at 2099, for a discussion of subranging ADC's. See also FIGS. 8 and 9, and discussion below.

In one embodiment, 30 coarse amplifiers, 30 coarse comparators, 19 fine amplifiers and 65 fine comparators are used.) The coarse amplifier A_(C) is connected to a capacitor C₁, which in turn is connected to either the output of a track-and-hold 101, or to V_(coarse) from the reference ladder 104. A two-phase clock, including phases φ₁ and φ₂, is used to control switches S₁, S₂ and S₃ of the coarse amplifier A_(C). When the phase φ₁ is on, the switches S₂ and S₃ are closed, the switch S₁ is open. With the switch S₃ closed, the coarse ADC amplifier A_(C) is in a reset mode, and the capacitor C₁ is connected to the reference ladder tap V_(coarse). Also on φ₁, the switch S₅ is closed, the switches S₄ and S₆ are open, and the fine capacitor C₂ is connected to an appropriate tap of the reference ladder V_(fine). Note that all of the switches as S₁-S₆ are typically field effect transistor (FET) switches (see FIG. 10, where the switches are S₁-S₆ illustrated as FET devices). The switch S₃ may be referred to as a coarse ADC reset switch, and the switch S₆ may be referred to as a fine ADC reset switch. When the phase φ₁ of the two-phase clock is on, the switches S₃ and S₂ are closed, the amplifier A_(C) is in a reset mode, and the left side of the capacitor C₁ is connected to a tap of the reference ladder (i.e., V_(coarse)). The switch S₁ is open when φ₁ is on.

On the opposite phase of the two-phase clock (φ₂), when φ₂ is high, the switch S₁ is closed, the switches S₂ and S₃ are open. The switches S₄ and S₆ are closed, and the fine amplifier A_(F) is in reset mode. Therefore the capacitor C₁ is connected to the track-and-hold output, and the amplifier A_(C) is in an amplify mode when the clock phase φ₂ is on.

Thus, the operation of the fine ADC amplifier A_(F) may be thought of as an inverse of the operation of the amplifier A_(C). In other words, when the clock phase φ₂ is on, the left side of the capacitor C₂ is connected to the track-and-hold through switch S₄, and the amplifier A_(F) is in the reset mode, since the switch S₆ is closed, and the switch S₅ is open. When the clock phase φ₁ is on, a switch S₅ is closed to connect the capacitor C₂ to V_(fine), (a subrange from the reference ladder 104), the switches S₄ and S₆ are open, and the amplifier A_(F) is in the amplify mode. The capacitors C₁ and C₂ are typically 50 to 200 femtofarads.

During the clock phase φ₁, the coarse amplifiers A_(C) are reset to the reference ladder 104, while the fine amplifiers A_(F) amplify the previous sample. During the clock phase φ₂, the coarse amplifiers A_(C) amplify the next sample, while the fine amplifiers A_(F) reset to the next sample.

Thus, there is no need to use a sample and hold amplifier, which uses one clock period for the operation of the coarse ADC 102, and one clock period for the operation of the fine ADC 105. With the arrangement shown in FIG. 2, the fine ADC amplifier A_(F) has a half cycle latency compared to the coarse ADC amplifier A_(C). Thus, one half of a clock cycle is available to do coarse quantization (and ½ cycle is available for fine quantization). During the reset phase, the auto-zero amplifiers can be connected either to the T/H or to a reference voltage. An important difference between these two possibilities is that resetting all amplifiers to a reference voltage requires half a clock cycle less latency in comparison with resetting to the T/H output voltage.

Therefore, if the coarse ADC amplifiers A_(C) are reset to a tap of the reference ladder 104, and the fine ADC amplifiers A_(F) are reset to the T/H 101, half a clock cycle now becomes available for performing the coarse quantization. (See also flowcharts in FIG. 6 and FIG. 7) Because a S/H is not needed with this timing approach, the resulting ADC can be more area- and power-efficient.

The coarse ADC amplifier A_(C) has one half of a clock cycle to set switches in the switch matrix 103, in order for the switch matrix 103 to pass the correct V_(fine) reference ladder 104 output to the fine ADC 105. While the amplifiers A_(C) and A_(F) require two phases to operate, the capacitors C₁ and C₂ subtract the V_(coarse) from the track-and-hold output, or V_(fine) from the track-and-hold output, respectively.

Phrased another way, there are two steps involved in the process:

1) Charge C₁, while the amplifier A_(C) is in a reset mode, and the amplifier A_(C) is providing a low impedance so that C₁ can be charged.

2) Release the reset, tie the capacitor C₁ to the amplifier A_(C) in order for it to amplify the track-and-hold output.

Thus, the track-and-hold 101 only outputs the signal for half a period, and C₁ is charged early, before the track-and-hold 101 is ready. When the track-and-hold 101 is ready, the amplifier A_(C) immediately does the amplification.

Although FIG. 2 shows only one coarse amplifier and one fine ADC amplifier A_(F), each can be part of an array of amplifiers in one embodiment of the invention. The amplifiers connect to the T/H during one half clock cycle, and to reference voltage taps (V_(coarse) or V_(fine)) from the reference ladder 104 during a second half clock cycle.

The amplifiers A_(C) and A_(F) are typically differential pair auto-zero amplifiers, with resistive load, preferably done in CMOS technology, e.g., NMOS or PMOS. Alternatively, the amplifiers A_(C) and A_(F) can be fabricated using bipolar technology.

FIG. 3 shows how a two-phase non-overlapping clock can be converted to a three phase clock to better fine tune the ability of the coarse and fine amplifiers A_(C), A_(F) and the comparators 107, 108 settle to their final values. As shown in FIG. 3 (and the flowcharts of FIG. 6 and 7), the phase φ₂ can be split up into φ₂ and φ_(2e), where φ_(2e) represents an early falling edge of the clock phase φ₂. Similarly, φ₁ can be used to generate a phase φ_(1d) _(—) _(fine), a delayed φ₁ phase of the clock used by the fine ADC 105. As illustrated by the notations at the top and bottom of FIG. 3, the coarse ADC amplifier A_(C) starts amplifying on the rising edge of φ_(2e), and stops amplifying on the falling edge φ_(2e). The fine ADC amplifier A_(F) is in a reset mode during that phase.

During the next phase of the clock, the coarse amplifier A_(C) begins resetting on the rising edge of φ₁, and stops resetting on the falling edge of φ₁. The comparators 107 of the coarse ADC 102 have from between the falling edge of φ_(2e) through the rising edge of φ_(1d) _(—) _(fine) to decide on whether they are latching 1 or 0 by comparing to a reference voltage from the reference ladder 104. (Each fine amplifier A_(F) is actually a cascade of amplifiers, GA, GB, GC, GD, as discussed below, and which is particularly useful in a pipelined architecture. The first amplifier in the cascade, GA, amplifies between the rising edge of φ_(1d) _(—) _(fine) through the falling edge of φ_(1d) _(—) _(fine).)

FIG. 4 further illustrates the operation of the amplifiers of the present invention in a situation where the fine ADC 105 has 4 cascaded stages (typically with a gain of 4× each), which are labeled GA, GB, GC and GD. (See FIG. 11, where two-cascaded stages are shown for both fine and coarse amplifiers A_(F) and A_(C), as one example.) In FIG. 4, the amplifier stage of the coarse ADC 102 is labeled GE, the coarse ADC comparator 107 is labeled CC, the fine ADC comparator 108 is labeled FC and the encoder is labeled ENC. The gray portions of FIG. 4 illustrate a progression of one sample's quantization down the amplifier cascade. First, the track-and-hold 101 is connected to the coarse ADC amplifier A_(C), during phase φ₂. Meanwhile, the coarse comparator 107 (CC) is reset during φ₂. The fine ADC amplifier A_(F) stage GA is also reset. During the next phase φ₁, the first stage GA of the fine ADC 105 amplifies, while the second stage GB resets. The process continues, as the signal moves in a pipelined manner down from GA to GB to GC to GD to the fine comparator 108 (FC), and ultimately to the encoder 106. The next quantization is directly behind the quantization just performed, moving from left to right in the figure, and offset by one clock cycle from the measurement illustrated in gray in FIG. 4.

FIGS. 5 and 6 illustrates the operation of the coarse/fine ADC's 102/105 in greater detail, showing the relationship between the clock phases of the three phase clock discussed above, and the operation of the various components shown in FIG. 2. As may be seen from FIGS. 5 and 6, the coarse capacitor C₁ is connected to the reference ladder 104 voltage V_(ref) (i.e., V_(coarse)) on the rising edge of the clock phase φ₁. The coarse amplifier 102 is also reset on the rising edge of φ₁, with the switch S₃ being closed until the falling edge of φ₁. The switch S₂ is kept closed until the falling edge of φ_(1d) (the delayed edge of φ₁).

On the rising edge of φ₂, the coarse capacitor C₁ is connected to the track-and-hold 101, and the coarse amplifier A_(C) begins amplifying the signal. The coarse comparator 107 (CC) is reset on the rising edge of φ₂, and the fine capacitor C₂ is connected to the track-and-hold voltage. The fine amplifier A_(F) is also reset on the rising edge of φ₂. The coarse capacitor C₁ is connected to the track-and-hold 101 until the falling edge of φ_(2d) (for delayed φ₂), the coarse comparator 107 (CC) begins latching at the falling edge of φ_(2e) (for early φ₂) and the fine amplifier A_(F) continues to be reset until the falling edge of φ₂. The fine capacitor C₂ is connected to the T/H through the delayed falling edge of φ₂ (φ_(2d))

On the next half clock cycle φ₁, the coarse comparator 107 (CC) is assumed to have latched at the rising edge of φ_(1d) _(—) _(fine), while the fine capacitor C₂ is connected to the reference voltage and fine amplifier A_(F) begins to amplify, also on the rising edge of φ_(1d) _(—) _(fine). The connection of the capacitor C₂ to the reference ladder 104 lasts until the falling edge of φ_(1d), and the fine amplifier A_(F) stops amplifying on the falling edge of φ_(1d).

The digital output of the coarse ADC tells the fine ADC 105 which subrange from the reference ladder voltage V_(ref) (i.e., V_(fine)) the switch matrix 103 should pass through to the fine ADC 105. Each amplifier amplifies only if there is a valid signal period. Here, the hold phase is the middle ⅓ phase of FIG. 5—both the coarse amplifier A_(C) and the fine amplifier A_(F) are looking at the signal. In a particular embodiment, an array of 30 coarse amplifiers and an array of 30 coarse comparators are used to get 31 subranges. The switch matrix 103 therefore connects to one out of 31 subranges. On the fine amplifier A_(F) side, the embodiment includes an array of 19 A-stage amplifiers, an array of 33 B-stage amplifiers, an array of 65 C-stage amplifiers and an array of 65 D-stage amplifiers, as well as an array of 65 fine comparators. The coarse ADC 102 of the present invention is illustrated in array form in FIG. 8, showing a plurality of amplifiers A_(C0)-A_(Cm), connected to a plurality of input capacitors C_(C0), and through the input capacitors C_(Cm) to a plurality of coarse taps from the reference ladder 104, taps V_(coarse,0) to V_(coarse,m). For 30 coarse amplifiers, m=29.

Similarly, the fine ADC 105 of the present invention is illustrated in array form in FIG. 9, showing a plurality of amplifiers A_(F0)-A_(Fn), connected to a plurality of input capacitors C_(F0), and through the input capacitors C_(Fn) to a plurality of fine taps from the reference ladder 104, taps V_(fine,)) to V_(fine,n). For 19 fine amplifiers, n=18. Note that although the auto-zero amplifiers A_(C0)-A_(Cm) and A_(F0)-A_(Fn) are shown as single ended in FIGS. 8 and 9, in actual application they are preferably differential amplifiers.

An “11 bit” output is actually converted to a 10-bit output, to compensate for conversion errors of the coarse ADC. With the approach of the present invention, there is no need to have to interleave ADC's running at ½ F_(sample). Here, a single ADC can be run at F_(sample), since there is delay of the latency of the fine ADC 105 by one half of a clock cycle.

Some timing refinements may be incorporated. The switches S₁, S₂ and S₄ shown on the left-side of the sampling capacitors C₁, C₂ in FIG. 2 can use slightly delayed clocks φ_(1d) and φ_(2d).

Furthermore, the coarse comparators 107 (CC) use an earlier clock signal φ_(1e), to give them somewhat more time to compare their input signal, thus improving their bit-error-rate. The switch S₅ connecting the fine amplifiers A_(F) to the reference ladder 104 use a delayed clock, φ_(1dfine), for the same reason. Basically, this implements a three-phase clock to operate the coarse ADC 102.

The proposed timing can be applied to all subranging ADC's to improve the required power and area.

It will be appreciated that the various aspects of the invention as further disclosed in related application Ser. No. 10/153,709, Filed: May 24, 2002, Titled: DISTRIBUTED AVERAGING ANALOG TO DIGITAL CONVERTER TOPOLOGY, Inventors: MULDER et al.; application Ser. No. 10/158,193, Filed: May 31, 2002; Titled: CLASS AB DIGITAL TO ANALOG CONVERTER/LINE DRIVER, Inventors: Jan MULDER et al.; application Ser. No. 10/158,595, Filed: May 31, 2002, Titled: HIGH SPEED ANALOG TO DIGITAL CONVERTER, Inventor: Jan MULDER; and application Ser. No. 10/158,774, Filed: May 31, 2002, Inventor: Jan MULDER; Titled: ANALOG TO DIGITAL CONVERTER WITH INTERPOLATION OF REFERENCE LADDER, Inventor: Jan MULDER, all of which are incorporated by reference herein, may be combined in various ways, or be integrated into a single integrated circuit or product.

It will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the invention as defined in the appended claims. Thus, the breadth and scope of the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents. 

What is claimed is:
 1. An analog to digital converter (ADC) comprising: a track-and-hold amplifier tracking an input voltage with its output; a coarse ADC amplifier connected to a coarse capacitor at its input and having a coarse ADC reset switch controlled by a first clock phase; a fine ADC amplifier connected to a fine capacitor at its input and having a fine ADC reset switch controlled by a second clock phase, wherein the track-and-hold amplifier is in a hold-mode during the second clock phase; a switch matrix that selects a set of reference voltages for use by the fine ADC amplifier based on an output of the coarse ADC amplifier, wherein the coarse capacitor is charged to a coarse reference voltage during the first clock phase and connected to the track-and-hold output voltage during the second clock phase, and wherein the fine capacitor is connected to a fine reference voltage during the first clock phase and charged to the track-and-hold output voltage during the second clock phase; and an encoder that converts outputs of the coarse and fine ADC amplifiers to an N-bit output.
 2. The analog to digital converter of claim 1, wherein the coarse ADC reset switch is a field effect transistor (FET).
 3. The analog to digital converter of claim 1, wherein the first and second clock phases are non-overlapping.
 4. The analog to digital converter of claim 1, wherein the fine ADC amplifier includes a plurality of cascaded amplifier stages.
 5. The analog to digital converter of claim 1, wherein the coarse ADC amplifier includes a plurality of cascaded amplifier stages.
 6. The analog to digital converter of claim 1, wherein the coarse capacitor is connected to the track-and-hold amplifier output on a delayed second phase.
 7. The analog to digital converter of claim 1, wherein the fine ADC capacitor is connected to the track-and-hold amplifier output on a delayed second clock phase and to the fine reference voltage during a delayed first clock phase.
 8. The analog to digital converter of claim 1, further including a switch that connects an output of the track-and-hold-amplifier to the coarse capacitor on the second clock phase.
 9. The analog to digital converter of claim 1, further including a coarse comparator that latches the output of the coarse ADC amplifier and outputs it to the encoder.
 10. The analog to digital converter of claim 1, further including a fine comparator that latches the output of the fine ADC amplifier and outputs it to the encoder.
 11. An analog to digital converter comprising: a track-and-hold amplifier tracking an input voltage; a first plurality of amplifiers each connected to a corresponding capacitor at its input, wherein the amplifiers of the first plurality are reset on a clock phase φ₁ and their corresponding capacitors are connected to an output of the track-and-hold on a clock phase φ₂, and wherein the track-and-hold amplifier is in a hold-mode on the clock phase φ₂; a second plurality of amplifiers each connected to a corresponding capacitor at its input, wherein the amplifiers of the second plurality are reset on the clock phase φ₂ and their corresponding capacitors are charged to the track-and-hold amplifier output voltage on the clock phase φ₂; a switch matrix that selects a set of reference voltages based on outputs of the first plurality of amplifiers, for input to the second plurality of amplifiers on the clock phase φ₁; and an encoder that converts outputs of the first and second pluralities of amplifiers to an N-bit output.
 12. The analog to digital converter of claim 11, further including FET switches that reset the first plurality of amplifiers on the clock phase φ₁.
 13. The analog to digital converter of claim 11, wherein the clock phases φ₁ and φ₂ are non-overlapping.
 14. The analog to digital converter of claim 11, wherein each of the second plurality of amplifiers includes a plurality of cascaded amplifier stages.
 15. The analog to digital converter of claim 11, wherein each of the first plurality of amplifiers includes a plurality of cascaded amplifier stages.
 16. The analog to digital converter of claim 11, wherein the capacitors of the first plurality of amplifiers are connected to the track-and-hold amplifier output on a delayed clock phase φ₂.
 17. The analog to digital converter of claim 11, wherein the capacitors of the second plurality of amplifiers are connected to the track-and-hold amplifier output on a delayed clock phase φ₂, and to the set of reference voltages on a delayed clock phase φ₁.
 18. The analog to digital converter of claim 11, further including switches that connect an output of the track-and-hold to the capacitors of the first plurality of amplifiers on the clock phase φ₂.
 19. The analog to digital converter of claim 11, further including a first plurality of comparators that latch the outputs of the first plurality of amplifiers and output them to the encoder.
 20. The analog to digital converter of claim 19, further including a second plurality of comparators that latch the outputs of the second plurality of amplifiers and output them to the encoder.
 21. An analog to digital converter comprising: a track-and-hold amplifier tracking an input voltage; a first capacitor connected to the track-and-hold amplifier on φ₂ and to a first reference voltage on a clock phase φ₁; a first amplifier that resets on the clock phase φ₁ and amplifies a voltage on the first capacitor on a clock phase φ₂; a first comparator for latching an output of the first amplifier on a clock phase φ_(1+1 cycle); a second capacitor connected to the track-and-hold on the clock phase φ₂ and to a second reference voltage on the clock phase φ₁, the second reference voltage selected based on the output of the first amplifier, wherein the track-and-hold amplifier is in a hold-mode on the clock phase φ₂; a second amplifier including a plurality of cascaded amplifier stages, wherein a first cascaded amplifier stage resets on the clock phase φ₂ and amplifies a voltage on the second capacitor on a clock phase φ_(1+1cycle), a second cascaded amplifier stage resets on the clock phase φ_(1+1cycle) and amplifies the voltage on the second capacitor on a clock phase φ_(2+1 cycle), a third cascaded amplifier stage resets on the clock phase φ_(2+1cycle) and amplifies the voltage on the second capacitor on a clock phase φ_(1+2 cycles), and so on; a second comparator for latching an output of a last cascaded amplifier stage; and an encoder converting outputs of the first and second comparators to an N-bit output.
 22. The analog to digital converter of claim 21, wherein the clock phases φ₁ and φ₂ are non-overlapping.
 23. The analog to digital converter of claim 21, wherein the first capacitor connects to the track-and-hold on a delayed clock phase φ₂.
 24. The analog to digital converter of claim 21, wherein the second capacitor is connected to the track-and-hold on a delayed clock phase φ₂ and to the second reference voltage on a delayed clock phase φ₁.
 25. The analog to digital converter of claim 21, further including a FET switch that connects the track-and-hold amplifier to the first capacitor on the clock phase φ₂.
 26. An analog to digital converter comprising: a track-and-hold amplifier tracking an input voltage; a first plurality of capacitors connected to an output of the track-and-hold on a clock phase φ₂ and to a first plurality of reference voltages on a clock phase φ₁; a first plurality of amplifiers that reset on the clock phase φ₁ and amplify voltages on the first plurality of capacitors on the clock phase φ₂; a first plurality of comparators for latching outputs of the first plurality of amplifiers; a second plurality of capacitors connected to the output of track-and-hold amplifier on the clock phase φ₂ and connected to a second plurality of reference voltages on the clock phase φ₁, the second plurality of reference voltages selected based on the outputs of the first plurality of amplifiers, wherein the track-and-hold amplifier is in a hold-mode on the clock phase φ₂; a second plurality of amplifiers, each including a plurality of cascaded amplifier stages, wherein the cascaded amplifier stages reset and amplify on alternating clock phases φ₁ and φ₂, wherein amplifiers of the first stage are reset on the clock phase φ₂ and amplify voltages of the second plurality of capacitors on the clock phase φ₁; a second plurality of comparators for latching outputs of a last amplifier stage; and an encoder converting outputs of the first and second pluralities of comparators to an N-bit output.
 27. The analog to digital converter of claim 26, wherein the clock phases φ₁ and φ₂ are non-overlapping.
 28. The analog to digital converter of claim 26, wherein the second plurality of capacitors are connected to the output of the track-and-hold amplifier during a delayed clock phase φ₂ and to the second plurality of reference voltages on a delayed clock phase φ₁.
 29. The analog to digital converter of claim 26, further including a plurality of switches that connect the output of the track-and-hold to the first plurality of capacitors on the clock phase φ₂.
 30. An analog to digital converter comprising: a track-and-hold amplifier tracking an input voltage; a first amplifier that resets on a clock phase φ₁ and amplifies a difference of an output of the track-and-hold amplifier and a first voltage reference on a clock phase φ₂, wherein the track-and-hold amplifier is in a hold-mode on the clock phase φ₂; a second amplifier that resets on the clock phase φ₂ and amplifies a difference of the output of the track-and-hold amplifier and a second reference voltage on the clock phase φ₁; a switch matrix that selects a first set of reference voltages for use by the second amplifier based on an output of the first amplifier; and an encoder that converts outputs of the first and second amplifiers to an N-bit output.
 31. A method of converting an analog voltage to a digital voltage comprising the steps of: resetting a first amplifier on a first clock phase; charging a first capacitor to a first reference voltage during the first clock phase; connecting the first capacitor to an input voltage during a second clock phase, wherein a track-and-hold amplifier is in a hold-mode during the second clock phase; selecting a second reference voltage based on an output of the first amplifier; connecting a second capacitor to the second reference voltage during the first clock phase; charging the second capacitor to the input voltage during the second clock phase; amplifying a voltage on the first capacitor on the second clock phase; resetting a second amplifier on the second clock phase; amplifying a voltage of the second capacitor on the first clock phase; and converting outputs of the first and second amplifiers to an N-bit output.
 32. A method of converting an analog voltage to a digital voltage comprising the steps of: on a first clock phase, charging a first capacitor to a first reference voltage, connecting a second capacitor to a second reference voltage, resetting a first amplifier, and amplifying a voltage of the second capacitor; on a second clock phase, connecting the first capacitor to an output of a track-and-hold amplifier, setting the track-and-hold amplifier to a hold mode, charging the second capacitor to the input voltage, resetting a second amplifier, selecting a set of reference voltages for use by the second amplifier based on an output of the first amplifier; and converting outputs of the first and second amplifiers to an N-bit output. 